In radio communication systems of next generation mobile communication, it is crucial to implement high speed data transmission. As a technique for implementing such high speed data transmission, a MIMO (Multiple Input Multiple Output) multiplexing system that simultaneously transmits signals at the same frequency from a plurality of transmission antennas and that demodulates (separates) the signals using a plurality of receive antennas, is attracting general attention.
FIG. 5 is a diagram of an arrangement of a typical MIMO transceiver device provided with an M-number of transmission antennas and an N-number of receive antennas, where M and N each denote an integer greater than or equal to 1. Referring to FIG. 5, the transmitting side includes transmission antennas 1-1 to 1-M and a transmitting apparatus 2, whereas the receiving side includes receive antennas 3-1 to 3-N and a receiving apparatus 4. The transmission antennas 1-1 to 1-M transmit respective different signals using the same frequency at the same time, and the receive antennas 3-1 to 3-N receive the signals, whereby high speed data transmission proportionate to the number of transmission antennas may be made without increasing the transmission bandwidth. The receiving side has to perform signal separation by demodulating signals received by the receive antennas 3-1 to 3-N into signals from the multiple transmission antennas 1-1 to 1-M.
Among a variety of methods for demodulating MIMO multiplexed signal, a method of linear filter reception is used as a simpler method.
In case the MIMO multiplexing scheme is used for the single carrier signal, in addition to interferences from other transmission antennas, multipaths of a desired transmission antenna signal become interferences, and signal reception via a filter that simultaneously suppresses these interferences is effective. There has been proposed a frequency equalizer that performs this processing by signal processing in the frequency domain to appreciably reduce the computational load (See, for example, Non-Patent Document 1).
Since the frequency equalizer calculates equalizing weights, channel estimation in the frequency domain becomes necessary. There has been proposed a method of directly transforming a reference received signal into a frequency domain and correlation of the received signal with a reference signal is taken in the frequency domain to estimate channel estimation (See, for example, Non-Patent Document 2).
FIG. 6 is a diagram showing a configuration, as related technique, of using a frequency domain equalizer and frequency domain channel estimation as described in Non-Patent Documents 1 and 2 for a MIMO receiving apparatus of a single carrier signal. The following explanation is made of a MIMO receiving apparatus of the related technique, shown in FIG. 6, with the number of the transmission antennas of M and with the number of the receive antennas of N, where M and N are each an integer greater than or equal to 1. In FIG. 6, an internal configuration of a receiving block 100-1, associated with the first receive antenna, is shown. Receiving blocks 100-1 to 100-N are all of the same configuration. In the notation of the reference numerals, as used in the present specification, -1 and -N at the trailing ends of respective reference numerals denote that the components in question belong to the first block and the N'th block, respectively.
The MIMO receiving apparatus includes the following components:
cyclic prefix (CP) removing units 101-1 to 101-N;
fast Fourier transform (FFT) units 102-1 to 102-N;
subcarrier demapping units 103-1 to 103-N;
reference signal generating units 104-1 to 104-M;
correlation processing units 105-1-1 to 105-M-N;
IFFT (Inverse Fast Fourier Transform) units 106-1-1 to 106-M-N;
noise path removing units 107-1-1 to 107-M-N;
FFT units 108-1-1 to 108-M-N;
a weight calculating unit 109:
an equalization filter 110; and
IDFT (Inverse Discrete Fourier Transform) units 111-1 to 111-M.
The CP removing units 101-1 to 101-N each input a received signal and removes a portion corresponding to CP from the received signal.
The FFT units 102-1 to 102-N input the received signal from which CP is removed by the CP removing units 101-1 to 101-N and each perform NFET point FFT (where NFFT is powers of 2) to output a received signal transformed into a frequency domain.
The subcarrier demapping units 103-1 to 103-N each input the received signal transformed into the frequency domain by the FFT units 102-1 to 102-N, select only the subcarrier of a desired user and decimate unneeded subcarriers.
The reference signal generating units 104-1 to 104-M generate reference signals used for processing for correlation with a reference received signal.
The reference signal generating units 104-1 to 104-M make use of                a ZF (Zero-Forcing) method that entirely cancels out a code characteristics of the reference received signal;        a MMSE (Minimum Mean Square Error) method that suppresses noise increase in the correlation processing; and        a clipping method.        
The correlation processing units 105-1-1 to 105-M-N each estimate a channel estimation value in the frequency domain based on correlation processing of the reference received signals and the reference signals. A channel estimation value HBF,m,n(k) of a transmission antenna m (1<=m<=M) and a receive antenna n (1<=n<=N) for a subcarrier k (1<=k<=NDFT) may be calculated by the following equation (1):HBF,m,n(k)=RRS,n(k)Xm′(k)  (1)
where Xm(k) denotes a reference signal of the transmission antenna in for the subcarrier k generated by a relevant one of the reference signal generating units 104-1 to 104-M and
RRS,n(k) denotes a reference received signal by the receive antenna n for the subcarrier k as obtained at a relevant one of subcarrier demapping units 103-1 to 103-N.
A suffix * denotes complex conjugate.
The IFFT units 106-1-1 to 106-M-N transform the channel estimation values in the frequency domain, estimated by the correlation processing units 105-1-1 to 105-M-N, into the channel response in the time domain.
The noise path removing units 107-1-1 to 107-M-N substitute zeros (0s) for signals (noise paths) at points including only noise to remove these noise-only point signals (noise paths) from the channel response which is an output of each of the IFFT units 106-1-1 to 106-M-N.
For each of the noise path removing units 107-1-1 to 107-M-N, a time window filter or noise threshold value control is used.
In the time window filter, it is supposed that the channel response is within a CP length, and signals of points other than the interval corresponding to the CP length are treated as a noise path and substituted with zeros (0s).
The noise threshold value control substitutes signals of points less than or equal to a preset threshold value with zeros (0s), as noise paths.
In case the time window filter and the noise threshold value control are used in conjunction, an average value of the noise outside the window of the time window filter may be used as a noise threshold value.
The FFT units 108-1-1 to 108-M-N each perform fast Fourier transform of the channel response from which the noise paths are removed by the noise path removing units 107-1-1 to 107-M-N, to output noise-suppressed channel estimation values in the frequency domain.
The weight calculating unit 109 inputs the noise-suppressed channel estimation values of the frequency domain, output from the FFT units 108-1-1 to 108-M-N, to calculate equalizing weights.
The weight calculating unit 109 generally uses an MMSE method or a ZF method.
The MMSE equalizing weight vector Wm(k) of the transmission antenna m for the subcarrier k is calculated by the following equation (2):
                                          W            m                    ⁡                      (            k            )                          =                                                            H                                  AF                  ,                  m                                H                            ⁡                              (                k                )                                      ⁡                          [                                                                    ∑                                                                  m                        ′                                            =                      1                                        M                                    ⁢                                                                                    H                                                  AF                          ,                                                      m                            ′                                                                                              ⁡                                              (                        k                        )                                                              ⁢                                                                  H                                                  AF                          ,                                                      m                            ′                                                                          H                                            ⁡                                              (                        k                        )                                                                                            +                                                      σ                    2                                    ⁢                  I                                            ]                                            -            1                                              (        2        )            
where HH denotes a complex conjugate transpose of H;
σ2 denotes a noise power; and
I denotes a unit matrix.
HAF,m(k) is an estimated channel vector between a transmission antenna m and a receive antenna for the subcarrier k. The estimated channel vector HAF,m(k) and the equalizing weight vector Wm(k) are defined respectively by the equations (3) and (4):HAF,m(k)=[HAF,m,1(k), HAF,m,2(k), . . . , HAF,m,N(k)]T  (3)Wm(k)=[Wm,1(k), Wm,2(k), . . . , Wm,N(k)]T  (4)
where T denotes transpose, and
each element of the estimated channel vector HAF,m(k) denotes a channel estimation value of the noise-suppressed frequency domain as an output of each of the FFT units 108-1-1 to 108-M-N.
The equalization filter 110 inputs the equalizing weights, calculated by the weight calculating unit 109, and the received signals, obtained by the subcarrier demapping units 103-1 to 103-N, to equalize the received signals in the frequency domain.
A transmission signal vector Y(k) for the subcarrier k, equalized and subjected to signal separation by the equalization filter 110 is calculated as shown in the equation (5), and defined by the equation (6):Y(k)=W(k)RD(k)  (5)Y(k)=[Y1(k), Y2(k), . . . , Ym(k)]T  (6)
where W(k) denotes a matrix of equalizing weights for the subcarrier k, and
RD(k) denotes a received signal vector for the subcarrier k. W(k) and RD(k) are respectively defined by equations (7) and (8), as given hereinbelow, respectively.
The elements of the received signal vector RD(k) denote received signals in the frequency domain as obtained by the subcarrier demapping units 103-1 to 103-N.W(k)=[W1(k), W2(k), . . . , WM(k)]T  (7)RD(k)=[RD,1(k), RD,2(k), . . . , RD,N(k)]T  (8)
The inverse discrete Fourier transform (IDFT) units 111-1 to 111-M input equalized frequency domain signals, output by the equalization filter 110, to perform MDR-points IDFT, where NIDFT is an integer greater than or equal to 2, thereby transforming the signals into time domain signals, which are output as demodulated signals.    Non-Patent Document 1:
Xu Zhu and Ross D. Murch, “Novel Frequency-Domain Equalization Architectures for a Single-Carrier Wireless MIMO System,”, IEEE VTC2002-Fall, pp. 874-878, September 2002.    Non-Patent Document 2:
Kimata and Yoshida, “A Study of Frequency Domain Demodulation Scheme in Uplink Single-Carrier IFDMA”, 2006 Shingku Sodai, B-5-36.